1. Technical Field
The present invention relates to communication systems and, more particularly, to analog-to-digital converters used within transceivers.
2. Related Art
Communication systems are known to support wireless and wire lined communications between wireless and/or wire lined communication devices. Such communication systems range from national and/or international cellular telephone systems to the Internet to point-to-point in-home wireless networks. Each type of communication system is constructed, and hence operates, in accordance with one or more communication standard. For instance, wireless communication systems may operate in accordance with one or more standards, including, but not limited to, IEEE 802.11, Bluetooth, advanced mobile phone services (AMPS), digital AMPS, global system for mobile communications (GSM), code division multiple access (CDMA), local multi-point distribution systems (LMDS), multi-channel-multi-point distribution service (MMDS), and/or variations thereof.
Depending on the type of wireless communication system, a wireless communication device, such as a cellular telephone, two-way radio, personal digital assistant (PDA), personal computer (PC), laptop computer, home entertainment equipment, etc., communicates directly or indirectly with other wireless communication devices. For direct communications (also known as point-to-point communications), the participating wireless communication devices tune their receivers and transmitters to the same channel or channels (e.g., one of a plurality of radio frequency (RF) carriers of the wireless communication system) and communicate over that channel(s). For indirect wireless communications, each wireless communication device communicates directly with an associated base station (e.g., for cellular services) and/or an associated access point (e.g., for an in-home or in-building wireless network) via an assigned channel. To complete a communication connection between the wireless communication devices, the associated base stations and/or associated access points communicate with each other directly, via a system controller, via the public switched telephone network (PSTN), via the Internet, and/or via some other wide area network.
Each wireless communication device includes a built-in radio transceiver (i.e., receiver and transmitter) or is coupled to an associated radio transceiver (e.g., a station for in-home and/or in-building wireless communication networks, RF modem, etc.) that performs analog signal processing tasks as a part of converting data to a radio frequency (RF) signal for transmission and a received RF signal to data.
As is known, the transmitter includes a data modulation stage, one or more intermediate frequency stages, and a power amplifier. The data modulation stage converts raw data into baseband signals in accordance with the particular wireless communication standard. The one or more intermediate frequency stages mix the baseband signals with one or more local oscillations to produce RF signals. The power amplifier amplifies the RF signals prior to transmission via an antenna.
As is also known, the receiver is coupled to the antenna and includes a low noise amplifier, one or more intermediate frequency stages, a filtering stage, and a data recovery stage. The low noise amplifier receives an inbound RF signal via the antenna and amplifies it. The one or more intermediate frequency stages mix the amplified RF signal with one or more local oscillations to convert the amplified RF signal into a baseband signal or an intermediate frequency (IF) signal. As used herein, the term “low IF” refers to both baseband and intermediate frequency signals.
A filtering stage filters the low IF signals to attenuate unwanted out of band signals to produce a filtered signal. The data recovery stage recovers raw data from the filtered signal in accordance with the particular wireless communication standard. Alternate designs being pursued at this time further include direct conversion radios that produce a direct frequency conversion often in a plurality of mixing steps or stages.
As an additional aspect, these designs are being pursued as a part of a drive to continually reduce circuit size and power consumption. Along these lines, such designs are being pursued with CMOS technology thereby presenting problems not addressed by prior art designs. For example, one common design goal is to provide an entire system on a single chip. The drive towards systems-on-chip solutions for wireless applications continues to replace traditional analog signal processing tasks with digital processing to exploit the continued shrinkage of digital CMOS technology.
One approach is to reduce analog signal processing performance requirements and to compensate for the relaxed performance requirements in the digital domain to provide required system performance. This approach is beneficial in that, in addition to the reduced silicon area requirements, the digital processing is insensitive to process and temperature variations. Applications for which this trend is observed include RF receivers where the received signal is digitized as early as possible in the receiver chain using a high dynamic range analog-to-digital converter (ADC), and in a variety of calibration circuits of the radio where signal levels must be measured accurately over a wide range of values. This trend thus increases the demand for embedded low-power, low-voltage ADCs providing high dynamic range in the interface between the analog and digital processing.
A class of ADCs capable of providing high dynamic range and particularly suitable for low-power and low-voltage implementation is known as continuous-time delta sigma analog-to-digital converters (CTΔΣADCs). These ADCs can be designed to operate with supply voltages in the range of 1.2V–1.5V and current consumption as low as a few hundred μAs.
FIG. 1 shows an example top-level block diagram of the simplest CTΔΣADC, namely, the first-order low pass CTΔΣADC. The input signal to the CTΔΣADC of FIG. 1 is a voltage source labeled s(t). An operational amplifier with negative capacitive feedback constitutes an integrator formed by the operational amplifier and capacitor in a feedback loop, which integrates the input current labeled is(t) flowing from an input signal s(t) to produce an analog integrator output voltage. A coarse (in this example 2-bit) quantizer converts the analog integrator output voltage signal to a digital format shown as y(t). The quantizer, by providing a 2-bit output, defines which of four voltage levels most closely match the analog integrator output voltage. More specifically, the quantizer produces a 2-bit output having values of 00, 01, 10 and 11.
The quantizer typically includes an array of comparators, essentially 1-bit ADCs, whose output is either “high” or “low” depending upon the magnitude of the integrator voltage relative to a reference signal generated by a reference generator. A digital-to-analog converter (DAC) provides a feedback current responsive to a logic value (“1” or “0”) of ADC output to the integrator. FIG. 2 shows one implementation of the 2-bit quantizer that produces the 2-bit feedback to the DAC. The quantizer sums the output values of the array of comparators to produce the 2-bit output discussed above.
FIG. 3 shows an alternative model of the first-order CTΔΣADC of FIG. 1, wherein the quantizer has been replaced with an additive noise source q(t). The model of FIG. 3 is one that represents the CTΔΣADC of FIG. 1. Because the operation of the quantizer is deterministic, a signal q(t) may be defined such that the CTΔΣADC of FIG. 3 behaves similarly to the CTΔΣADC of FIG. 1. The digital ADC output, here denoted y(t), can then be written as a sum of two terms, namely, a term related to the input signal, ys(t), and a term related to the quantization noise, yq(t), i.e.,y(t)=ys(t)+yq(t).  (1)
By employing feedback around the integrator and quantizer combination, it is possible to suppress the quantization noise component yq(t) in a limited frequency range around DC. Specifically, it can be shown that yq(t) results from q(t) being filtered by a first-order high-pass filter, commonly referred to as the noise transfer function, NTF(s), i.e., in terms of Laplace transforms,Yq(s)=NTF(s)×Q(s).  (2)
Similarly, for a low-frequency input signal s(t), it can be shown that the signal component ys(t) equals the input signal, i.e., in terms of Laplace transforms,Ys(s)=S(s).  (3)
The above properties explain the terminology “low pass” CTΔΣADC; if s(t) is a low frequency input signal, the ADC output y(t) closely resembles s(t) when considering only the low frequency region of y(t), i.e., the ADC “passes” signals of low frequency from analog to digital format without alteration. Furthermore, the low pass CTΔΣADC of FIG. 1 is of first-order since the single integrator gives rise to a first order high pass filter. More integrators can, in principle, be added to yield higher order filtering of the quantization noise as is described further below. Generally, an Nth order CTΔΣADC contains N integrators.
Ideally, in equation (2), the quantization noise q(t) is uncorrelated with the input signal s(t) and closely resembles white noise of power Δ2/12, where Δ is the quantizer step size (see FIG. 2) as long as the input signal is limited such that the quantizer operates in the no-overload region. In this case, the two terms that constitute y(t) in equation (1) are uncorrelated, or, equivalently, yq(t) closely resembles white noise, uncorrelated with the input, and filtered by the high pass filter NTF(s). In this case, since NTF(s) is deterministic, the power of the quantization noise measured over a given signal band-width, fc, of the ADC output y(t) can be determined using standard linear systems analysis as
                              P          n                =                              ∫                          f              =              0                                      f              =                              f                c                                              ⁢                                                    Δ                2                            12                        ⁢                                                                            N                  ⁢                                                                          ⁢                  T                  ⁢                                                                          ⁢                                      F                    ⁡                                          (                                              ⅇ                                                  j2                          ⁢                                                                                                          ⁢                          π                          ⁢                                                                                                          ⁢                          f                                                                    )                                                                                                  2                        ⁢                                                  ⁢                                          ⅆ                f                            .                                                          (        4        )            
For a given known input signal power, Ps, the signal-to-noise ratio (SNR)—a measure of the quality of the analog-to-digital conversion process—can then be calculated a-priori according to
                              S          ⁢                                          ⁢          N          ⁢                                          ⁢          R                =                                            P              s                                      P              n                                .                                    (        5        )            
Some properties of the ideal CTΔΣADC, where q(t) resembles white and random noise, follow from (4) and (5). For a given fixed fc, which depends upon the particular application, the SNR depends upon the input as would be expected from a linear system with q(t) contributing constant noise power at the output. In other words, any change of signal power leads to an identical change of SNR in the ADC output. For example, suppose that the signal power is doubled, e.g., increases by 6 dB, it then follows from (5) that the SNR increases by 6 dB.
Having a high SNR of the analog-to-digital converted signal is extremely important in almost all applications. Typically, the SNR delivered by the ADC must be within tight tolerances to satisfy strict performance and behavior requirements of the overall system. For example, SNR's less than 85 dB are undesirable in many applications such as RF receivers for cellular telephony including GSM and EDGE systems. In zero-IF and 100 kHz IF CTΔΣADCs, a digital filter is commonly used to achieve such a high SNR by filtering out frequency components above fc in the ADC output signal. However, existing approaches to achieving an SNR of 85 dB at zero-IF and 100 kHz IF cannot accommodate for an IF of 200 kHz while maintaining 85 dB SNR over the signal bandwidth. This is due to the fact that traditional CTΔΣADCs employ integrators, and are thus inherently designed for lowpass function. CTΔΣADCs that operate at 200 kHz IF provide several advantages over zero-IF or 100 kHz IF CTΔΣADCs, such as decreased receiver sensitivity to circuit flicker noise and circuit DC offset. Thus, a need exists for a CTΔΣADC architecture that optimizes the signal-to-noise ratio while operating at a low IF of 200 kHz.